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MC33260
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GreenLineTM Compact Power Factor Controller: Innovative Circuit for Cost Effective Solutions
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The MC33260 is a controller for Power Factor Correction preconverters meeting international standard requirements in electronic ballast and off-line power conversion applications. Designed to drive a free frequency discontinuous mode, it can also be synchronized and in any case, it features very effective protections that ensure a safe and reliable operation. This circuit is also optimized to offer extremely compact and cost effective PFC solutions. While it requires a minimum number of external components, the MC33260 can control the follower boost operation that is an innovative mode allowing a drastic size reduction of both the inductor and the power switch. Ultimately, the solution system cost is significantly lowered. Also able to function in a traditional way (constant output voltage regulation level), any intermediary solutions can be easily implemented. This flexibility makes it ideal to optimally cope with a wide range of applications. Standard Constant Output Voltage or "Follower Boost" Mode Switch Mode Operation: Voltage Mode Latching PWM for Cycle-by-Cycle On-Time Control Constant On-Time Operation That Saves the Use of an Extra Multiplier Totem Pole Output Gate Drive Undervoltage Lockout with Hysteresis Low Start-Up and Operating Current Improved Regulation Block Dynamic Behavior Synchronization Capability Internally Trimmed Reference Current Source Overvoltage Protection: Output Overvoltage Detection Undervoltage Protection: Protection Against Open Loop Effective Zero Current Detection Accurate and Adjustable Maximum On-Time Limitation Overcurrent Protection ESD Protection on Each Pin
TYPICAL APPLICATION
D1...D4 Filtering Capacitor L1 VCC M1 Ro Sync D1 + C1 LOAD (SMPS, Lamp Ballast,...)
8 1 DIP-8 P SUFFIX CASE 626
PIN CONNECTIONS AND MARKING DIAGRAM
Feedback Input Vcontrol Oscillator Capacitor (CT) Current Sense Input 1 MC33260 AWL YYWW (Top View) 2 3 4 8 VCC 7 Gate Drive 6 Gnd 5 Synchronization Input
General Features
* * * * * * * * * * * * * * * *
AWL = Manufacturing Code YYWW = Date Code
ORDERING INFORMATION
Device MC33260P Package Plastic DIP-8 Shipping 50 Units / Rail
Safety Features
Vcontrol R OCP CT
1 2 3 4
8 7 6 5
R cs
This document contains information on a product under development. ON Semiconductor reserves the right to change or discontinue this product without notice.
(c) Semiconductor Components Industries, LLC, 1999
MC33260
1
November, 1999 - Rev. 1
Publication Order Number: MC33260/D
MC33260
BLOCK DIAGRAM
Vo
Current Mirror IOSC - ch = CT 3 11 V 1 0 15 pF 97%Iref Output_Ctrl Iref Vref 1.5 V Io Vreg 300 k Vcontrol 2 2 x IO x IO Iref Io Io Io Io Iref Current Mirror 1 FB
IovpH/IovpL REGULATOR Enable Vref + Iref r r - + Ics (205 mA) r -60 mV 1 Current Sense 4 11 V Output_Ctrl ThStdwn LEB 0 + - 11 V/8.5 V Iuvp UVP - OVP
11 V
+ -
Synchro 5 11 V
Synchro Arrangement VCC 8 Drive 7 Gnd S R Q PWM Latch Q MC33260 Output_Ctrl 6
+ - PWM Comparator
R R
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MC33260
MAXIMUM RATINGS
Rating Gate Drive Current (Pin 7)* Source Sink VCC (Pin 8) Maximum Voltage Input Voltage Power Dissipation and Thermal Characteristics P Suffix, DIP Package Maximum Power Dissipation @ TA = 85C Thermal Resistance Junction to Air Operating Junction Temperature Operating Ambient Temperature *The maximum package power dissipation must be observed. Pin # 7 IO(Source) IO(Sink) 8 (Vcc)max Vin -500 500 16 -0.3 to +10 V V Symbol Value Unit mA
PD RJA TJ TA
600 100 150 -40 to +105
mW C/W C C
ELECTRICAL CHARACTERISTICS (VCC = 13 V, TJ = 25C for typical values, TJ = -40 to 105C for min/max values
unless otherwise noted.) Characteristic GATE DRIVE SECTION Gate Drive Resistor Source Resistor @ Ipin7 = 100 mA Sink Resistor @ Ipin7 = 100 mA Gate Drive Voltage Rise Time (From 3 V Up to 9 V) (Note 1) Output Voltage Falling Time (From 9 V Down to 3 V) (Note 1) OSCILLATOR SECTION Maximum Oscillator Swing Charge Current @ Ipin1 = 100 A Charge Current @ Ipin1 = 200 A Ratio Multiplier Gain Over Maximum Swing @ Ipin1=100 A Ratio Multiplier Gain Over Maximum Swing @ Ipin1=200 A Average Internal Pin 3 Capacitance Over Oscillator Maximum Swing (Vpin3 Varying From 0 Up to 1.5 V) (Note 2) Discharge Time (CT = 1 nF) REGULATION SECTION Regulation High Current Reference Ratio (Regulation Low Current Reference) / Ireg-H Pin 2 Impedance Pin 1 Clamp Voltage @ Ipin1 = 100 A Pin 1 Clamp Voltage @ Ipin1 = 200 A 1 1 1 1 1 Ireg-H Ireg-L / Ireg-H Zpin3 Vpin1-100 Vpin1-200 192 0.965 -- 1.5 2 200 0.97 300 2.1 2.6 208 0.98 -- 2.5 3 A -- k V V 3 3 3 3 3 3 VT Icharge Icharge Kosc Kosc Cint 1.4 87.5 350 5600 5600 10 1.5 100 400 6400 6400 15 1.6 112.5 450 7200 7200 20 V A A 1/(V.A) 1/(V.A) pF 7 ROL ROH 7 7 tr tf 10 5 -- -- 20 10 50 50 35 25 -- -- ns ns Pin # Symbol Min Typ Max Unit
3
Tdisch
--
0.5
1
s
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MC33260
ELECTRICAL CHARACTERISTICS (VCC = 13 V, TJ = 25C for typical values, TJ = -40 to 105C for min/max values
unless otherwise noted.) Characteristic CURRENT SENSE SECTION Zero Current Detection Comparator Threshold Negative Clamp Level (Ipin2 = -1 mA) Bias Current @ Vpin4 = VZCD-th Propagation Delay (Vpin4 > VZCD-th) to Gate Drive High Pin 4 Internal Current Source Leading Edge Blanking Duration OverCurrent Protection Propagation Delay (Pin 4 < VZCD-th to Gate Drive Low) SYNCHRONIZATION SECTION Synchronization Threshold Negative Clamp Level (Ipin5 = -1 mA) Minimum Off-Time Minimum Required Synchronization Pulse Duration OVERVOLTAGE PROTECTION SECTION OverVoltage Protection High Current Threshold and Ireg-H Difference OverVoltage Protection Low Current Threshold and Ireg-H Difference Ratio (IOVP-H/IOVP-L) Propagation Delay (Ipin1 > 110% Iref to Gate Drive Low) UNDERVOLTAGE PROTECTION SECTION Ratio (UnderVoltage Protection Current Threshold) / Ireg-H Propagation Delay (Ipin1 < 12% Iref to Gate Drive Low) THERMAL SHUTDOWN SECTION Thermal Shutdown Threshold Hysteresis VCC UNDERVOLTAGE LOCKOUT SECTION Start-Up Threshold Disable Voltage After Threshold Turn-On TOTAL DEVICE Power Supply Current Start-Up (VCC = 5 V with VCC Increasing) Operating @ Ipin1 = 200 A 8 ICC -- -- 0.1 4 0.25 8 mA 8 8 Vstup-th Vdisable 9.7 7.4 11 8.5 12.3 9.6 V V 7 7 Tstdwn Tstdwn -- -- 150 30 -- -- C C 1 7 IUVP/Ireg-H TUVP 12 -- 14 500 16 -- % ns 1 1 1 7 IOVP-H-Ireg-H IOVP-L-Ireg-H IOVP-H / IOVP-L TOVP 8 0 1.02 -- 13 -- -- 500 18 -- -- -- A -- -- ns 5 5 7 5 Vsync-th Cl-neg Toff Tsync 0.8 -- 1.5 -- 1 -0.7 2.1 -- 1.2 -- 2.7 0.5 V V s s 7 4 4 4 7 4 VZCD-th Cl-neg Ib-cs TZCD IOCP LEB TOCP -90 -- -0.2 -- 192 -- 100 -60 -0.7 -- 500 205 400 160 -30 -- -- -- 218 -- 240 mV V A ns A ns ns Pin # Symbol Min Typ Max Unit
NOTES: (1) 1 nF being connected between the pin 7 and ground. (2) Guaranteed by design. (3) No load is connected to the gate drive which is kept high during the test.
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MC33260
Vcontrol : REGULATION BLOCK OUTPUT (V) Vcontrol : REGULATION BLOCK OUTPUT (V) 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 0 20 40 60 80 100 120 140 160 180 200 220 240 Ipin1: FEEDBACK CURRENT (A) - 40C 25C 105C 1.6 1.4 1.2 1.0 0.8 0.6 0.4 0.2 0 185 190 195 200 205 210 Ipin1: FEEDBACK CURRENT (A) - 40C 25C 105C
Figure 1. Regulation Block Output versus Feedback Current
Figure 2. Regulation Block Output versus Feedback Current
1.340 MAXIMUM OSCILLATOR SWING (V) FEEDBACK INPUT VOLTAGE (V) 1.335 1.330 1.325 1.320 1.315 1.310 1.305 1.300 -40 -20 0 20 40 60 80 100
3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 0 20 40 60 80 100 120 140 160 180 200 220 240 JUNCTION TEMPERATURE (C) Ipin1: FEEDBACK CURRENT (A) - 40C 25C 105C
Figure 3. Maximum Oscillator Swing versus Temperature
Figure 4. Feedback Input Voltage versus Feedback Current
I osc-ch , OSCILLATOR CHARGE CURRENT (m A)
500 450 400 350 300 250 200 150 100 50 0 0 20 40 60 80 100 120 140 160 180 200 220 240 Ipin1: FEEDBACK CURRENT (A) - 40C 25C 105C
I osc-ch , OSCILLATOR CHARGE CURRENT (m A)
410 Ipin1 = 200 mA 405 400 395 390 385 -40
-20
0
20
40
60
80
100
JUNCTION TEMPERATURE (C)
Figure 5. Oscillator Charge Current versus Feedback Current
Figure 6. Oscillator Charge Current versus Temperature
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MC33260
104 OSCILLATOR CHARGE CURRENT ( A) 103 102 101 100 99 98 97 -40 -20 0 20 40 60 80 100 ON-TIME ( s) Ipin1 = 100 mA 120 100 80 60 40 20 0 30 50 70 90 110 130 150 170 190 210 TJ, JUNCTION TEMPERATURE (C) Ipin1: FEEDBACK CURRENT (mA) - 40C 25C 105C 1 nF Connected to Pin 3
Figure 7. Oscillator Charge Current versus Temperature
REGULATION AND CS CURRENT SOURCE ( A)
Figure 8. On-Time versus Feedback Current
75 65 ON-TIME ( s) 55 45 35 25 15 50 60 70 80 90 100 Ipin1: FEEDBACK CURRENT (A) - 40C 25C 105C 1 nF Connected to Pin 3
207 206 205 204 203 202 201 200 199 198 197 -40 IregH IOCP
-20
0
20
40
60
80
100
TJ, JUNCTION TEMPERATURE (C)
Figure 9. On-Time versus Feedback Current
Figure 10. Internal Current Sources versus Temperature
1.07 (IovpH /I ref ), (I ovpL /I ref ), (I regL /I ref ) 1.05 1.04 1.03 1.02 1.01 1.00 0.99 0.98 0.97 0.96 -40 (IregL/Iref) (IovpL/Iref) UNDERVOLTAGE RATIO (I uvp /I ref ) 40 60 80 100 1.06 (IovpH/Iref)
0.150 0.148 0.146 0.144 0.142 0.140 0.138 0.136 0.134 0.132 0.130 -40
-20
0
20
-20
0
20
40
60
80
100
TJ, JUNCTION TEMPERATURE (C)
TJ, JUNCTION TEMPERATURE (C)
Figure 11. (IovpH/Iref), (IovpL/Iref), (IregL/Iref) versus Temperature
Figure 12. Undervoltage Ratio versus Temperature
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MC33260
-54.8 CURRENT SENSE THRESHOLD (mV) I CC , CIRCUIT CONSUMPTION (mA) -55 -55.2 -55.4 -55.6 -55.8 -56 -56.2 -56.4 -56.6 -40 -20 0 20 40 60 80 100 4.5 4.0 3.5 3.0 2.5 2.0 1.5 1.0 0.5 0 0 2 4 6 8 10 12 14 16 VCC: SUPPLY VOLTAGE (V) - 40C 25C 105C
TJ, JUNCTION TEMPERATURE (C)
Figure 13. Current Sense Threshold versus Temperature
Figure 14. Circuit Consumption versus Supply Voltage
OSCILLATOR PIN INTERNAL CAPACITANCE (pF)
20
Vgate -40C 25C 1 25C VCC = 12 V Cgate = 1 nF
15
10 Icross-cond (50 mA/div) 5 105C 2 0 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 Ch1 10.0 V Ch2 10.0 mVW Vcontrol: PIN 2 VOLTAGE (V) M 1.00 ms Ch1 600 mV
Figure 15. Oscillator Pin Internal Capacitance
Figure 16. Gate Drive Cross Conduction
Vgate - 40C VCC = 12 V Cgate = 1 nF
Vgate 105C VCC = 12 V Cgate = 1 nF
1
1
Icross-cond (50 mA/div)
Icross-cond (50 mA/div)
2 Ch1 10.0 V Ch2 10.0 mVW M 1.00 ms Ch1 600 mV
2 Ch1 10.0 V Ch2 10.0 mVW M 1.00 ms Ch1 600 mV
Figure 17. Gate Drive Cross Conduction
Figure 18. Gate Drive Cross Conduction
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MC33260
PIN FUNCTION DESCRIPTION
Pin No. 1 Function Feedback Input Description This pin is designed to receive a current that is proportional to the preconverter output voltage. This information is used for both the regulation and the overvoltage and undervoltage protections. The current drawn by this pin is internally squared to be used as oscillator capacitor charge current. This pin makes available the regulation block output. The capacitor connected between this pin and ground, adjusts the control bandwidth. It is typically set below 20 Hz to obtain a nondistorted input current. The circuit uses an on-time control mode. This on-time is controlled by comparing the CT voltage to the Vcontrol voltage. CT is charged by the squared feedback current. This pin is designed to receive a negative voltage signal proportional to the current flowing through the inductor. This information is generally built using a sense resistor. The Zero Current Detection prevents any restart as long as the pin 4 voltage is below (-60 mV). This pin is also used to perform the peak current limitation. The overcurrent threshold is programmed by the resistor connected between the pin and the external current sense resistor. This pin is designed to receive a synchronization signal. For instance, it enables to synchronize the PFC preconverter to the associated SMPS. If not used, this pin must be grounded. This pin must be connected to the preregulator ground. The gate drive current capability is suited to drive an IGBT or a power MOSFET. This pin is the positive supply of the IC. The circuit turns on when VCC becomes higher than 11 V, the operating range after start-up being 8.5 V up to 16 V.
2
Vcontrol
3 4
Oscillator Capacitor (CT) Zero Current Detection Input
5 6 7 8
Synchronization Input Ground Gate Drive VCC
APPLICATION SCHEMATIC
D1...D4
Filtering Capacitor L1 D1 + C1 1 2 Vcontrol 3 4 ROCP CT MC33260 8 7 6 5 Sync VCC M1 Ro Load (SMPS, Lamp Ballast,...)
Rcs
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MC33260
FUNCTIONAL DESCRIPTION
INTRODUCTION OPERATION DESCRIPTION
The need of meeting the requirements of legislation on line current harmonic content, results in an increasing demand for cost effective solutions to comply with the Power Factor regulations. This data sheet describes a monolithic controller specially designed for this purpose. Most off-line appliances use a bridge rectifier associated to a huge bulk capacitor to derive raw dc voltage from the utility ac line.
Rectifiers AC Line Converter
+
Bulk Storage Capacitor
Load
Figure 19. Typical Circuit Without PFC
This technique results in a high harmonic content and in poor power factor ratios. In effect, the simple rectification technique draws power from the mains when the instantaneous ac voltage exceeds the capacitor voltage. This occurs near the line voltage peak and results in a high charge current spike. Consequently, a poor power factor (in the range of 0.5 - 0.7) is generated, resulting in an apparent input power that is much higher than the real power.
Vpk Rectified DC 0 Line Sag AC Line Voltage 0 AC Line Current
The MC33260 is optimized to just as well drive a free running as a synchronized discontinuous voltage mode. It also features valuable protections (overvoltage and undervoltage protection, overcurrent limitation, ...) that make the PFC preregulator very safe and reliable while requiring very few external components. In particular, it is able to safely face any uncontrolled direct charges of the output capacitor from the mains which occur when the output voltage is lower than the input voltage (start-up, overload, ...). In addition to the low count of elements, the circuit can control an innovative mode named "Follower Boost" that permits to significantly reduce the size of the preconverter inductor and power MOSFET. With this technique, the output regulation level is not forced to a constant value, but can vary according to the a.c. line amplitude and to the power. The gap between the output voltage and the ac line is then lowered, what allows the preconverter inductor and power MOSFET size reduction. Finally, this method brings a significant cost reduction. A description of the functional blocks is given below.
REGULATION SECTION
Connecting a resistor between the output voltage to be regulated and the pin 1, a feedback current is obtained. Typically, this current is built by connecting a resistor between the output voltage and the pin 1. Its value is then given by the following equation:
I
+ pin1
Vo
* Vpin1
Ro
Figure 20. Line Waveforms Without PFC
Active solutions are the most popular way to meet the legislation requirements. They consist of inserting a PFC pre-regulator between the rectifier bridge and the bulk capacitor. This interface is, in fact, a step-up SMPS that outputs a constant voltage while drawing a sinusoidal current from the line.
Rectifiers AC Line PFC Preconverter High Frequency Bypass Capacitor Bulk Storage Capacitor Converter
where: Ro is the feedback resistor, Vo is the output voltage, Vpin1 is the pin 1 clamp value. The feedback current is compared to the reference current so that the regulation block outputs a signal following the characteristic depicted in Figure 22. According to the power and the input voltage, the output voltage regulation level varies between two values (Vo)regL and (Vo)regH corresponding to the IregL and IregH levels.
Regulation Block Output 1.5 V
MC33260
Load
+
Io Ireg-L (97%Iref) Ireg-H (Iref)
Figure 21. PFC Preconverter
Figure 22. Regulation Characteristic
The MC33260 was developed to control an active solution with the goal of increasing its robustness while lowering its global cost.
The feedback resistor must be chosen so that the feedback current should equal the internal current source IregH when
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MC33260
the output voltage exceeds the chosen upper regulation voltage [(Vo)regH]. Consequently:
Ro 2 Vo R2 o
+
Vo
V regH pin1 I regH
*
I
charge
+
* Vpin1
I ref
2
In practice, Vpin1 is small compared to (Vo)regH and this equation can be simplified as follows (IregH being also replaced by its typical value 200 A):
Ro
[5
Vo
regH

The regulation block output is connected to the pin 2 through a 300 k resistor. The pin 2 voltage (Vcontrol) is compared to the oscillator sawtooth for PWM control. An external capacitor must be connected between pin 2 and ground, for external loop compensation. The bandwidth is typically set below 20 Hz so that the regulation block output should be relatively constant over a given ac line cycle. This integration that results in a constant on-time over the ac line period, prevents the mains frequency output ripple from distorting the ac line current.
OSCILLATOR SECTION
where: Vo is the output voltage, Ro is the feedback resistor, Vpin1 is the pin 1 clamp voltage. In practice, Vpin1 that is in the range of 2.5 V, is very small compared to Vo. The equation can then be simplified by neglecting Vpin1:
I
[ R22 charge
o
V2 o I
ref
It must be noticed that the oscillator terminal (pin 3) has an internal capacitance (Cint) that varies versus the pin 3 voltage. Over the oscillator swing, its average value typically equals 15 pF (min 10 pF, max 20 pF). The total oscillator capacitor is then the sum of the internal and external capacitors.
C pin3
+ CT ) Cint
The oscillator consists of three phases: * Charge Phase: The oscillator capacitor voltage grows up linearly from its bottom value (ground) until it exceeds Vcontrol (regulation block output voltage). At that moment, the PWM latch output gets low and the oscillator discharge sequence is set. * Discharge Phase: The oscillator capacitor is abruptly discharged down to its valley value (0 V). * Waiting Phase: At the end of the discharge sequence, the oscillator voltage is maintained in a low state until the PWM latch is set again.
Icharge = 2
PWM LATCH SECTION
The MC33260 operates in voltage mode: the regulation block output (Vcontrol - pin 2 voltage) is compared to the oscillator sawtooth so that the gate drive signal (pin 7) is high until the oscillator ramp exceeds Vcontrol. The on-time is then given by the following equation:
t on
+
C
pin3 I
V ch
control
Io Io / Iref
Output_Ctrl
1 CT 3
0
where: ton is the on-time, Cpin3 is the total oscillator capacitor (sum of the internal and external capacitor), Icharge is the oscillator charge current (pin 3 current), Vcontrol is the pin 2 voltage (regulation block output). Consequently, replacing Icharge by the expression given in the Oscillator Section:
t on
1 15 pF
0
+
R2 o
I
ref
pin3 2 V2 o
C
V
control
Figure 23. Oscillator
The oscillator charge current is dependent on the feedback current (Io). In effect
I charge
One can notice that the on-time depends on Vo (preconverter output voltage) and that the on-time is maximum when Vcontrol is maximum (1.5 V typically). At a given Vo, the maximum on-time is then expressed by the following equation:
t on max
+2
I2 o I ref
+
C
pin3
R2 o
I 2
ref V2 o
V
control max
where: Icharge is the oscillator charge current, Io is the feedback current (drawn by pin 1), Iref is the internal reference current (200 A). So, the oscillator charge current is linked to the output voltage level as follows:
This equation can be simplified replacing
2 / [(Vcontrol)max * Iref] by Kosc
Refer to Electrical Characteristics, Oscillator Section. Then:
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MC33260
t on max
+ Kpin3
osc
C
R2 o V2 o
Zero Current Detection
This equation shows that the maximum on-time is inversely proportional to the squared output voltage. This property is used for follower boost operation (refer to Follower Boost section).
CURRENT SENSE BLOCK
The inductor current is converted into a voltage by inserting a ground referenced resistor (Rcs) in series with the input diodes bridge (and the input filtering capacitor). Therefore a negative voltage proportional to the inductor current is built:
V cs
The Zero Current Detection function guarantees that the MOSFET cannot turn on as long as the inductor current hasn't reached zero (discontinuous mode). The pin 4 voltage is simply compared to the (-60 mV) threshold so that as long as Vcs is lower than this threshold, the circuit gate drive signal is kept in low state. Consequently, no power MOSFET turn on is possible until the inductor current is measured as smaller than (60 mV/Rcs) that is, the inductor current nearly equals zero.
Iocp (205 mA) D1...D4 1 ROCP Rcs VOCP 4 LEB 0 Output_Ctrl -60 mV + - S PWM Latch Output_Ctrl R Q R
+*
R cs
I
L
where: IL is the inductor current, Rcs is the current sense resistor, Vcs is the measured Rcs voltage.
Inductor Current Power Switch Drive
To Output Buffer (Output_Ctrl Low <=> Gate Drive in Low State)
Figure 25. Current Sense Block
Time
Overcurrent Protection
Rcs Voltage
During the power switch conduction (i.e. when the Gate Drive pin voltage is high), a current source is applied to the pin 4. A voltage drop VOCP is then generated across the resistor ROCP that is connected between the sense resistor and the Current Sense pin (refer to Figure 25). So, instead of Vcs, the sum (Vcs + VOCP) is compared to (-60 mV) and the maximum permissible current is the solution of the following equation:
* Rcs
Ipk max
) VOCP + * 60 mV ) 60
Pin 4 Voltage
VOCP
where: Ipkmax is maximum allowed current, Rcs is the sensing resistor. The overcurrent threshold is then:
Ipk max
-60 mV Zero Current Detection VOCP = ROCP IOCP An overcurrent is detected if Vpin4 crosses the threshold (-60 mV) during the Power Switch on state
+
R
OCP
I
OCP R cs
10
*3
Figure 24. Current Sensing
The negative signal Vcs is applied to the current sense through a resistor ROCP. The pin is internally protected by a negative clamp (-0.7 V) that prevents substrate injection. As long as the pin 4 voltage is lower than (-60 mV), the Current Sense comparator resets the PWM latch to force the gate drive signal low state. In that condition, the power MOSFET cannot be on. During the on-time, the pin 4 information is used for the overcurrent limitation while it serves the zero current detection during the off time.
where: ROCP is the resistor connected between the pin and the sensing resistor (Rcs), IOCP is the current supplied by the Current Sense pin when the gate drive signal is high (power switch conduction phase). IOCP equals 205 A typically. Practically, the VOCP offset is high compared to 60 mV and the precedent equation can be simplified. The maximum current is then given by the following equation:
Ipk max
[ ROCP R cs
0.205
Consequently, the ROCP resistor can program the OCP level whatever the Rcs value is. This gives a high freedom in the choice of Rcs. In particular, the inrush resistor can be utilized.
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MC33260
Th-Stdwn VCC
5 Current Sense Comparator - +
Synchronization Arrangement OVP, UVP
S Q PWM Latch
Output Buffer
7
ZCD & OCP R & Q
-60 mV PWM Latch Comparator
Output_Ctrl
+ Oscillator Sawtooth
-
Vcontrol (Vpin2 - Regulation Output)
Figure 26. PWM Latch
A LEB (Leading Edge Blanking) has been implemented. This circuitry disconnects the Current Sense comparator from pin 4 and disables it during the 400 first ns of the power switch conduction. This prevents the block from reacting on the current spikes that generally occur at power switch turn on. Consequently, proper operation does not require any filtering capacitor on pin 4.
PROTECTIONS OCP (Overcurrent Protection)
Practically, Vpin1 that is in the range of 2.5 V, can be neglected. The equation can then be simplified:
V ovpH
+ Ro
I
ovpH

On the other hand, the OVP low threshold is:
V ovpL
+ Vpin1 )
Ro
I
ovpL
where Iovp-L is the internal low OVP current threshold. Consequently, Vpin1 being neglected:
V ovpL
Refer to Current Sense Block.
OVP (Overvoltage Protection)
+ Ro
I
ovpL
The feedback current (Io) is compared to a threshold current (IovpH). If it exceeds this value, the gate drive signal is maintained low until this current gets lower than a second level (IovpL).
Gate Drive Enable Vcontrol
The OVP hysteresis prevents erratic behavior. IovpL is guaranteed to be higher than IregH (refer to parameters specification). This ensures that the OVP function doesn't interfere with the regulation one.
UVP (Undervoltage Protection)
Io Iuvp IregL IregH IovpL IovpH
This function detects when the feedback current is lower than 14% of Iref. In this case, the PWM latch is reset and the power switch is kept off. This protection is useful to: * Protect the preregulator from working in too low mains conditions. * To detect the feedback current absence (in case of a nonproper connection for instance). The UVP threshold is:
V uvp
[ Vpin1 )
R o
I uvp (V)
Figure 27. Internal Current Thresholds
Practically (Vpin1 being neglected),
V uvp
So, the OVP upper threshold is:
V ovpH
+ Vpin1 )
+ Ro
I uvp
Ro
I
ovpH
Maximum On-Time Limitation
where: Ro is the feedback resistor that is connected between pin 1 and the output voltage, Iovp-H is the internal upper OVP current threshold, Vpin1 is the pin 1 clamp voltage.
As explained in PWM Latch, the maximum on-time is accurately controlled.
Pin Protection
All the pins are ESD protected.
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In particular, a 11 V zener diode is internally connected between the terminal and ground on the following pins: Feedback, Vcontrol, Oscillator, Current Sense, and Synchronization.
Sync 5 1V Rsync 2 ms + S1 - UVLO R2 & PWM Latch Set Q1 Q1 High <=> Synchronization Mode
S2 1V R2 Output_Ctrl Q2
Figure 28. Synchronization Arrangement SYNCHRONIZATION BLOCK
The MC33260 features two modes of operation: * Free Running Discontinuous Mode: The power switch is turned on as soon as there is no current left in the inductor (Zero Current Detection). This mode is simply obtained by grounding the synchronization terminal (pin 5). * Synchronization Mode: This mode is set as soon as a signal crossing the 1 V threshold, is applied to the pin 5. In this case, operation in free running can only be recovered after a new circuit start-up. In this mode, the power switch cannot turn on before the two following conditions are fulfilled. -- Still, the zero current must have been detected. -- The precedent turn on must have been followed by (at least) one synchronization raising edge crossing the 1 V threshold. In other words, the synchronization acts to prolong the power switch off time. Consequently, a proper synchronized operation requires that the current cycle (on-time + inductor demagnetization) is shorter than the synchronization period. Practically, the inductor must be chosen accordingly. Otherwise, the system will keep working in free running discontinuous mode. Figure 33 illustrates this behavior. It must be noticed that whatever the mode is, a 2 s minimum off-time is forced. This delay limits the switching frequency in light load conditions.
OUTPUT SECTION
have been controlled to typically equal 50 ns while loaded by 1 nF.
REFERENCE SECTION
An internal reference current source (Iref) is trimmed to be 4% accurate over the temperature range (the typical value is 200 A). Iref is the reference used for the regulation (IregH = Iref).
UNDERVOLTAGE LOCKOUT SECTION
An Undervoltage Lockout comparator has been implemented to guarantee that the integrated circuit is operating only if its supply voltage (VCC) is high enough to enable a proper working. The UVLO comparator monitors the pin 8 voltage and when it exceeds 11 V, the device gets active. To prevent erratic operation as the threshold is crossed, 2.5 V of hysteresis is provided. The circuit off state consumption is very low: in the range of 100 A @ VCC = 5 V. This consumption varies versus VCC as the circuit presents a resistive load in this mode.
THERMAL SHUTDOWN
An internal thermal circuitry is provided to disable the circuit gate drive and then to prevent it from oscillating, if the junction temperature exceeds 150C typically. The output stage is again enabled when the temperature drops below 120C typically (30C hysteresis).
FOLLOWER BOOST
The output stage contains a totem pole optimized to minimize the cross conduction current during high speed operation. The gate drive is kept in a sinking mode whenever the Undervoltage Lockout is active. The rise and fall times
Traditional PFC preconverters provide the load with a fixed and regulated voltage that generally equals 230 V or 400 V according to the mains type (U.S., European, or universal). In the "Follower Boost" operation, the preconverter output regulation level is not fixed but varies linearly versus the ac line amplitude at a given input power.
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MC33260
Traditional Output
traditional preconverter follower boost preconverter
IL
Ipk
Vo (Follower Boost) Vac
Vin Vin time
Vin
Load
IL
Vin
Vout IL
Figure 29. Follower Boost Characteristics
the power switch is on
the power switch is off
This technique aims at reducing the gap between the output and the input voltages to minimize the boost efficiency degradation.
Follower Boost Benefits
Figure 30. Off-Time Duration Increase
Follower Boost Implementation
The boost presents two phases: * The on-time during which the power switch is on. The inductor current grows up linearly according to a slope (Vin/Lp), where Vin is the instantaneous input voltage and Lp the inductor value. * The off-time during which the power switch is off. The inductor current decreases linearly according the slope (Vo - Vin) / Lp, where Vo is the output voltage. This sequence that terminates when the current equals zero, has a duration that is inversely proportional to the gap between the output and input voltages. Consequently, the off-time duration becomes longer in follower boost. Consequently, for a given peak inductor current, the longer the off time, the smaller power switch duty cycle and then its conduction dissipation. This is the first benefit of this technique: the MOSFET on-time losses are reduced. The increase of the off time duration also results in a switching frequency diminution (for a given inductor value). Given that in practise, the boost inductor is selected big enough to limit the switching frequency down to an acceptable level, one can immediately see the second benefit of the follower boost: it allows the use of smaller, lighter and cheaper inductors compared to traditional systems. Finally, this technique utilization brings a drastic system cost reduction by lowering the size and then the cost of both the inductor and the power switch.
In the MC33260, the on-time is differently controlled according to the feedback current level. Two areas can be defined: * When the feedback current is higher than IregL (refer to regulation section), the regulation block output (Vcontrol) is modulated to force the output voltage to a desired value. * On the other hand, when the feedback current is lower than IregL, the regulation block output and therefore, the on-time are maximum. As explained in PWM Latch Section, the on-time is then inversely proportional to the output voltage square. The Follower Boost is active in these conditions in which the on-time is simply limited by the output voltage level. Note: In this equation, the feedback pin voltage (Vpin1) is neglected compared to the output voltage (refer to the PWM Latch Section).
t on
+
t on max
+ Kpin3
osc
C
R2 o V2 o
where: Cpin3 is the total oscillator capacitor (sum of the internal and external capacitors - Cint + CT), Kosc is the ratio (oscillator swing over oscillator gain), Vo is the output voltage, Ro is the feedback resistor.
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MC33260
On the other hand, the boost topology has its own rule that dictates the on-time necessary to deliver the required power:
t on
Vo (Pin)min Regulation Block is Active Vo = Vpk
+4
Lp V2 pk
P
in
where: Vpk is the peak ac line voltage, Lp is the inductor value, Pin is the input power. Combining the two equations, one can obtain the Follower Boost equation:
Vo
Pin (Pin)max
non usable area
+ R2o
C K osc
pin3 Lp
P
V in
pk
Vac VacLL Vac VacHL
Consequently, a linear dependency links the output voltage to the ac line amplitude at a given input power.
The Regulation Block is Active Output Voltage Input Power (Vac)max Vac Pin (Vac)min Vo ton = k/Vo2
Figure 32. Follower Boost Output Voltage Mode Selection
Output Voltage Input Power
ton
on-time
Figure 31. Follower Boost Characteristics
The behavior of the output voltage is depicted in Figures 31 and 32. In particular, Figure 31 illustrates how the output voltage converges to a stable equilibrium level. First, at a given ac line voltage, the on-time is dictated by the power demand. Then, the follower boost characteristic makes correspond one output voltage level to this on-time. Combining these two laws, it appears that the power level forces the output voltage. One can notice that the system is fully stable: * If an output voltage increase makes it move away from its equilibrium value, the on-time will immediately diminish according to the follower boost law. This will result in a delivered power decrease. Consequently, the supplied power being too low, the output voltage will decrease back, * In the same way, if the output voltage decreases, more power will be transferred and then the output voltage will increase back.
The operation mode is simply selected by adjusting the oscillator capacitor value. As shown in Figure 32, the output voltage first has an increasing linear characteristic versus the ac line magnitude and then is clamped down to the regulation value. In the traditional mode, the linear area must be rejected. This is achieved by dimensioning the oscillator capacitor so that the boost can deliver the maximum power while the output voltage equals its regulation level and this, whatever the given input voltage. Practically, that means that whatever the power and input voltage conditions are, the follower boost would generate output voltages values higher than the regulation level, if there was no regulation block. In other words, if (Vo)regL is the low output regulation level:
Vo regL
v R2o
4
C K osc
T
) Cint
P
V max
Lp
pk
in
Consequently,
C
T
w * Cint )
K osc
Lp R2 o
P
max in V2 pk
2 V o regL
Using IregL (regulation block current reference), this equation can be simplified as follows:
C
T
w * Cint )
4
K osc
Lp
P V2 pk
in
max
I2 regL
In the Follower Boost case, the oscillator capacitor must be chosen so that the wished characteristics are obtained. Consequently, the simple choice of the oscillator capacitor enables the mode selection.
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MC33260
Synchronization Signal
Zero Current Detection
2 ms Delay 2 ms 2 ms 2 ms 2 ms
Oscillator
Vcontrol
Circuit Output
205 mA
Ics
Inductor Current
1
2
3
4
case no. 1: the turn on is delayed by the Zero Current Detection cases no. 2 and no. 3: the turn on is delayed by the synchronization signal case no. 4: the turn on is delayed by the minimum off-time (2 ms)
Figure 33. Typical Waveforms http://onsemi.com
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MC33260
MAIN DESIGN EQUATIONS (Note 1)
rms Input Current (Iac) I ac
+h
Po Vac 2 (P o) max V acLL Po f ac Co Vo
2 V acLL
(preconverter efficiency) is generally in the range of 90 - 95%.
Maximum Inductor Peak Current ((Ipk)max): (I pk ) max
+2 + 2p
Vo 2
h
(Ipk)max is the maximum inductor current.
Output Voltage Peak to Peak 100Hz (120Hz) Ripple ((Vo)pk-pk): (DVo ) pk-pk Inductor Value (Lp): Lp 2 t Vo fac is the ac line frequency (50 or 60Hz)
+
* VacLL
(I pk
V acLL
) max
t is the maximum switching period. (t=40s) for universal mains operation and (t=20s) for narrow range are generally used. (Rds)on is the MOSFET drain source on-time resistor. In Follower Boost, the ratio (VacLL/Vo) is higher. The on-time MOSFET losses are then reduced The Average Diode Current depends on the power and on the output voltage. This formula indicates the required dissipation capability for Rcs (current sense resistor). The overcurrent threshold is adjusted by ROCP at a given Rcs. Rcs can be a preconverter inrush resistor
Maximum Power MOSFET Conduction Losses ((pon)max): (Pon ) max
[1 3
(Rds)on
(I
pk
) max 2
1
* 1.2
V acLL Vo
Maximum Average Diode Current (Id): (I ) max d Current Sense Resistor Losses (pRcs): pR cs
+ (Po))max (Vo min
(I pk ) 2 max
+1 6
(Rds)on
Over Current Protection Resistor (ROCP): R cs (I ) max pk R OCP 0.205
[
(kW)
Oscillator External Capacitor Value (CT): -Traditional Operation 2K C - Follower Boost: Vo Feedback Resistor (Ro): Ro
w * Cint ) T + R2o +
osc
Lp
(P ) max in V2 ac
I2 regL
C K osc
T
) Cint
Lp P in Vo [ 200
The Follower Boost characteristic is adjusted by the CT choice. Th T di i l Mode is l l d by The Traditional M d i also selected b CT. Cint is the oscillator pin internal capacitor.
V pk The output voltage regulation level is adjusted by Ro.
(Vo ) reg I
* Vpin1
(MW)
regH
Note 1. The preconverter design requires the following characteristics specification: - (Vo)reg: desired output voltage regulation level - (Vo)pk-pk: admissible output peak to peak ripple voltage - Po: desired output power - Vac: ac rms operating line voltage - VacLL: minimum ac rms operating line voltage
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MC33260
L1 1N4007 D1 D2 90 to 270 Vac EMI Filter C1 330 nF 500 Vdc R2 1 MW 0.25 W Q1 MTP4N50E 320 mH D5 MUR460E R1 1 MW 0.25 W + C2 47 mF 450 V
80 W Load (SMPS, Lamp Ballast,...)
D3
D4
R4 15 kW/0.25 W
R3 1 W/2 W
R5 22 W/0.25 W Feedback Block Vreg Vcontrol C3 2 680 nF Vreg 300 k 97%.Iref Iref Io Io 1.5 V Iref Vprot Regulation Block Io Io Iref UVP, OVP Vref Iref MC33260 - + 11 V/8.5 V VCC 8 (- - -) Iuvp IovpL IovpH Io Vprot ThStdwn Output Buffer Drive 7 Gnd R PWM Latch S Q Q 6
1
REGULATOR Enable
PWM Comp Oscillator
I osc-ch
+ - Iocp (205 mA) 0 1 1 0 -60 mV Current Sense Block + - Output 4 LEB
+
2x|0x|0 I ref
CT C4 330 pF 3 15 pF
Output
Synchronization Block
Synchro 5
L1: Coilcraft N2881 - A (primary: 62 turns of # 22 AWG - Secondary: 5 turns of # 22 AWG Core: Coilcraft PT2510, EE 25 L1: Gap: 0.072 total for a primary inductance (Lp) of 320 mH)
Figure 34. 80 W Wide Mains Power Factor Corrector POWER FACTOR CONTROLLER TEST DATA*
AC Line Input Current Harmonic Distortion (% Ifund) Vrms (V) 90 110 135 180 220 240 260 Pin (W) 88.2 86.3 85.2 87.0 84.7 85.3 84.0 PF (-) 0.991 0.996 0.995 0.994 0.982 0.975 0.967 Ifund (mA) 990 782 642 480 385 359 330 THD 8.1 7.0 8.2 9.5 15 16.5 18.8 H2 0.07 0.05 0.03 0.16 0.5 0.7 0.7 H3 5.9 2.7 1.5 4.0 8.4 9.0 11.0 H5 4.3 5.7 6.8 6.5 7.8 7.8 7.0 H7 1.5 1.1 1.1 3.1 5.3 7.4 9.0 H9 1.7 0.8 1.5 4.0 1.9 3.8 4.0 Vo (V) 181 222 265 360 379 384 392 Vo (V) 31.2 26.4 20.8 16.0 14.0 14.0 13.2 DC Output Io (mA) 440 360 300 225 210 210 205 Po (W) 79.6 79.9 79.5 81.0 79.6 80.6 80.4 (%) 90.2 92.6 93.3 93.1 94.4 94.5 95.7
*Measurements performed using Voltech PM1200 ac power analysis.
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MC33260
Rstup D1...D4 + 15 V Cpin8
r
1 MC33260 2 3 4
8 7 6 5
VCC
+
Figure 35. Circuit Supply Voltage MC33260 VCC SUPPLY VOLTAGE
In some applications, the arrangement shown in Figure 35 must be implemented to supply the circuit. A start-up resistor is connected between the rectified voltage (or one-half wave) to charge the MC33260 VCC up to its start-up threshold (11 V typically). The MC33260 turns on and the VCC capacitor (Cpin8) starts to be charged by the PFC transformer auxiliary winding. A resistor, r (in the range of 22 ) and a 15 V zener should be added to protect the circuit from excessive voltages.
When the PFC preconverter is loaded by an SMPS, the MC33260 should preferably be supplied by the SMPS itself. In this configuration, the SMPS starts first and the PFC gets active when the MC33260 VCC supplied by the power supply, exceeds the device start-up level. With this configuration, the PFC preconverter doesn't require any auxiliary winding and finally a simple coil can be used.
PCB LAYOUT
The connections of the oscillator and Vcontrol capacitors should be as short as possible.
Preconverter Output + +
1 2 3 4 MC33260
8 7 6 5
VCC + + +
+ +
SMPS Driver
Figure 36. Preconverter loaded by a Flyback SMPS: MC33260 VCC Supply
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MC33260
PACKAGE DIMENSIONS DIP-8 P SUFFIX PLASTIC PACKAGE CASE 626-05 ISSUE K
8
5
-B-
1 4
NOTES: 1. DIMENSION L TO CENTER OF LEAD WHEN FORMED PARALLEL. 2. PACKAGE CONTOUR OPTIONAL (ROUND OR SQUARE CORNERS). 3. DIMENSIONING AND TOLERANCING PER ANSI Y14.5M, 1982. DIM A B C D F G H J K L M N MILLIMETERS MIN MAX 9.40 10.16 6.10 6.60 3.94 4.45 0.38 0.51 1.02 1.78 2.54 BSC 0.76 1.27 0.20 0.30 2.92 3.43 7.62 BSC --- 10_ 0.76 1.01 INCHES MIN MAX 0.370 0.400 0.240 0.260 0.155 0.175 0.015 0.020 0.040 0.070 0.100 BSC 0.030 0.050 0.008 0.012 0.115 0.135 0.300 BSC --- 10_ 0.030 0.040
F
NOTE 2
-A- L
C -T-
SEATING PLANE
J N D K
M
M
H
G 0.13 (0.005) TA
M
B
M
STYLE 1: PIN 1. 2. 3. 4. 5. 6. 7. 8.
AC IN DC + IN DC - IN AC IN GROUND OUTPUT AUXILIARY VCC
ON Semiconductor and are trademarks of Semiconductor Components Industries, LLC (SCILLC). SCILLC reserves the right to make changes without further notice to any products herein. SCILLC makes no warranty, representation or guarantee regarding the suitability of its products for any particular purpose, nor does SCILLC assume any liability arising out of the application or use of any product or circuit, and specifically disclaims any and all liability, including without limitation special, consequential or incidental damages. "Typical" parameters which may be provided in SCILLC data sheets and/or specifications can and do vary in different applications and actual performance may vary over time. All operating parameters, including "Typicals" must be validated for each customer application by customer's technical experts. SCILLC does not convey any license under its patent rights nor the rights of others. SCILLC products are not designed, intended, or authorized for use as components in systems intended for surgical implant into the body, or other applications intended to support or sustain life, or for any other application in which the failure of the SCILLC product could create a situation where personal injury or death may occur. Should Buyer purchase or use SCILLC products for any such unintended or unauthorized application, Buyer shall indemnify and hold SCILLC and its officers, employees, subsidiaries, affiliates, and distributors harmless against all claims, costs, damages, and expenses, and reasonable attorney fees arising out of, directly or indirectly, any claim of personal injury or death associated with such unintended or unauthorized use, even if such claim alleges that SCILLC was negligent regarding the design or manufacture of the part. SCILLC is an Equal Opportunity/Affirmative Action Employer.
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MC33260/D

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